Power transfer device using an oscillator

ABSTRACT

A power transfer device includes an oscillator circuit having a first node, a second node, and a control terminal. The oscillator circuit includes a cascode circuit comprising transistors having a first conductivity type and a first breakdown voltage. The cascode circuit is coupled to the control terminal, the first node, and the second node. The oscillator circuit includes a latch circuit coupled between the cascode circuit and a first power supply node. The latch circuit includes cross-coupled transistors having the first conductivity type and a second breakdown voltage. The first breakdown voltage is greater than the second breakdown voltage. The oscillator circuit may be configured to develop a pseudo-differential signal on the first node and the second node. The pseudo-differential signal may have a peak voltage of at least three times a voltage level of an input DC signal on a second power supply node.

CROSS-REFERENCE TO RELATED APPLICATION(S)

This application is a continuation of U.S. patent application Ser. No.15/835,230, filed Dec. 7, 2017, entitled “Power Transfer Device Using anOscillator,” naming Mohammad Al-Shyoukh, Krishna Pentakota, and StefanA. Mastovich as inventors, which application is incorporated herein byreference.

BACKGROUND Field of the Invention

This invention relates to isolation technology and more particularly toproviding power across an isolation barrier.

Description of the Related Art

Referring to FIG. 1, a conventional high-power system (e.g., a systemhaving a power level greater than approximately 1 W) uses a powerconverter including standard transformer 109, e.g., a discretetransformer with a ferrite core and high efficiency to transfer poweracross the isolation barrier. Depending on the complexity of the drivecircuitry, the conventional high-power system may achieve power transferefficiencies of approximately 70%-approximately 95%. In order toregulate the output voltage, communications channel 104 provides anynecessary feedback signals across the isolation barrier. Although thestandard transformer implementation is efficient, the size and cost ofthe standard transformer implementation may be prohibitive for use insome applications. Thus, low-cost, isolated power transfer systemshaving high power transfer efficiency are desired.

SUMMARY OF EMBODIMENTS OF THE INVENTION

In at least one embodiment, a method for operating a power transferdevice includes converting an input DC signal into an output DC signal.The output DC signal is electrically isolated from the input DC signal.The converting includes biasing an oscillator circuit with the input DCsignal and developing a pseudo-differential signal on a first node ofthe oscillator circuit and a second node of the oscillator circuit. Thepseudo-differential signal has a peak voltage of at least three times aninput voltage level of the input DC signal.

In at least one embodiment, a method for manufacturing a power transferdevice includes forming a primary-side circuit on a first integratedcircuit die configured to receive an input DC signal. The methodincludes forming a secondary-side circuit on a second integrated circuitdie configured to provide an output DC signal. The method includesforming a transformer on an insulating substrate. The method includeselectrically coupling the transformer between the primary-side circuitand the secondary-side circuit. The method includes disposing theinsulating substrate directly on conductors in a package housing thefirst integrated circuit die and the second integrated circuit die. Thetransformer is electrically isolated from the conductors.

In at least one embodiment, a power transfer device has a primary sideand a secondary side isolated from the primary side by an isolationbarrier. The power transfer device includes a first power supply nodeand a second power supply node. The power transfer device includes anoscillator circuit coupled to the first power supply node and the secondpower supply node. The oscillator circuit includes a primary-sideconductive coil having a center tap coupled to the first power supplynode. The oscillator circuit is configured as a DC/AC power convertercircuit that receives, on the first power supply node, an input DCsignal, and regulates an output voltage of the secondary side based on afeedback signal received from the secondary side via a communicationchannel across the isolation barrier.

BRIEF DESCRIPTION OF THE DRAWINGS

The present invention may be better understood, and its numerousobjects, features, and advantages made apparent to those skilled in theart by referencing the accompanying drawings.

FIG. 1 illustrates a functional block diagram of a conventional circuitfor transferring power across an isolation barrier using feedback toregulate the output signal.

FIG. 2 illustrates a functional block diagram of a system fortransferring power across an isolation barrier using feedback toregulate the output signal.

FIG. 3 illustrates a circuit diagram of an exemplary oscillator circuit.

FIGS. 4-7 and 9 illustrate circuit diagrams of exemplary oscillatorcircuits including a latch circuit and a cascode circuit consistent withembodiments of the invention.

FIG. 8 illustrates a circuit diagram of an exemplary snubber circuit ofFIGS. 7 and 9 consistent with embodiments of the invention.

FIG. 10 illustrates exemplary signal waveforms of thepseudo-differential signal generated by oscillator circuits of FIGS. 4-7and 9 consistent with embodiments of the invention.

FIG. 11 illustrates exemplary signal waveforms for the isolated powertransfer system of FIG. 2.

FIGS. 12-14 illustrate circuit diagrams of exemplary rectifier circuitsconsistent with at least one embodiment of the invention.

FIG. 15 illustrates exemplary waveforms for half-cycle signals andassociated output signal of the rectifier circuit of FIG. 14.

FIG. 16 illustrates a functional block diagram of a system fortransferring power across an isolation barrier using feedback toregulate the output voltage consistent with at least one embodiment ofthe invention.

FIG. 17 illustrates a plan view of a packaged power transfer deviceconsistent with at least one embodiment of the invention.

FIG. 18 illustrates a cross-sectional view of an air-core transformer ofthe packaged power transfer device of FIG. 17 consistent with at leastone embodiment of the invention.

FIGS. 19 and 20 illustrate circuit diagrams of exemplary embodiments ofthe feedback circuit of FIG. 16 consistent with at least one embodimentof the invention.

FIG. 21 illustrates exemplary signal waveforms for the isolated outputvoltage and enable signal of FIG. 16.

The use of the same reference symbols in different drawings indicatessimilar or identical items.

DETAILED DESCRIPTION

A low-cost, power transfer device includes a transformer formed on aninsulating substrate disposed on conductive structures within anintegrated circuit package. A primary winding of the transformer iscoupled to a first integrated circuit to form a DC/AC power converterand a secondary winding of the transformer is coupled to a secondintegrated circuit to form an AC/DC power converter. The first andsecond integrated circuits are electrically isolated from each other,i.e., no current flows between the first and second integrated circuits.

Referring to FIG. 2, power transfer device 200 includes DC/AC powerconverter circuit 202, which uses input DC signal V_(DD1) to biasconductive coil 206 of transformer 209, and AC/DC power convertercircuit 204, which uses conductive coil 208 of transformer 209 to drivecapacitor C1. In at least one embodiment, an oscillator circuit includesconductive coil 206 and DC/AC power converter circuit 202 to form afast-starting oscillator stage configured to operate as a Class-D poweramplifier that is configured as a primary-side power converter stage.This primary-side power converter stage may be tuned to oscillate with aparticular frequency, e.g., approximately 60 MHz-400 MHz, using variablecapacitors. Power transfer device 200 regulates output voltage V_(DD2)by turning on and off DC/AC power converter circuit 202 using feedbackinformation received from feedback circuit 210 via capacitive channel220 to communicatively couple electrically isolated integrated circuits.Capacitive channel 220 communicates the feedback information across anisolation barrier from the secondary side to the primary side. Thefeedback may be provided as a digital signal communicated across theisolation barrier using transmitter 216 and receiver 214 that implementon-off keying (OOK) modulation of the information transmitted acrosscapacitive channel 220.

In at least one embodiment, the primary-side power converter stageformed by conductive coil 206 and the oscillator in DC/AC powerconverter circuit 202 operates as a high-efficiency Class-D poweramplifier. Class-D operation may cause a pseudo-differential signal onnodes TX+ and TX− to have peak voltage levels (e.g., 15 V) up to, orslightly greater than, 3.2×V_(DD1). Such voltage levels are nottolerated by conventional CMOS devices (e.g., conventional CMOStransistors operate up to 1.2×V_(DD1)). Conventional oscillator circuit302 of FIG. 3, which includes a latch circuit formed from conventionaln-type transistor 306 and n-type transistor 308, is inadequate tosupport the high gate-to-source voltage levels and drain-to-sourcevoltage levels of the target specifications for the primary-side powerconverter stage.

Referring to FIG. 4, in at least one embodiment, oscillator circuit 402includes a latch circuit formed by latch transistor 408 and latchtransistor 410, which are n-type transistors cross-coupled to each otherand coupled to cascode transistor 404 and cascode transistor 406, whichare also n-type transistors. Latch transistor 408 and latch transistor410 are on the primary side (e.g., low voltage side) of the powertransfer device and are configured to pump energy into the LC tankcircuit of oscillator circuit 402 at a frequency that is determined bypassive system elements. Conductive coil 206 (i.e., the primary-sidewinding of transformer 209) can experience voltages as high as 3×V_(DD1)due to the Class-D mode operation (e.g., the pseudo-differential signalon nodes V_(Ha) and V_(Hb) having voltage levels in a range between2.6×V_(DD1) and 3.2×V_(DD1)) of oscillator circuit 402. Oscillatorcircuit 402 is selectively enabled via cascode transistors 404 and 406,which can cut off the current path to transformer 209. The Class-Doperation of oscillator circuit 402 reduces transition times between theon (i.e., conducting) portion of oscillator circuit 402 and the off(i.e., non-conducting) portion of oscillator circuit 402, which realizesnear-instant-or near-zero voltage switching in the primary-side powerconverter stage, thereby increasing efficiency by limiting the timeduration in which both n-type transistors consume power and reducing oreliminating overshoots or undesired transients in the delivery of energyto the secondary-side power converter stage.

In at least one embodiment of oscillator circuit 402, cascode transistor404 and cascode transistor 406 are laterally-diffused drain metal oxidesemiconductor (LDMOS) transistors engineered for a high breakdownvoltage. An exemplary LDMOS transistor can sustain high drain-to-sourcevoltages (e.g., tens of Volts) while having low equivalenton-resistances (R_(dson)) in response to being driven into the linearmode of transistor operation. In at least one embodiment of the powertransfer device, transistor 404 and transistor 406 are 18 V LDMOS n-typetransistors, which are available in an exemplary manufacturing processfor mixed-signal integrated circuits (e.g., a bipolar-CMOS-DMOSmanufacturing process). Other transistors used by oscillator circuit 402(e.g., latch transistor 408 and latch transistor 410) are conventional 5V CMOS devices that have a breakdown voltage that is just over V_(DD1)(e.g., a breakdown voltage in a range greater than 5 V, but less than 6V). Cascode transistor 404 and cascode transistor 406 shield the latchcircuit from high voltages. The drain terminals of cascode transistor404 and cascode transistor 406 can support high drain-to-source voltageswings while corresponding gate-to-source voltages are maintained withinreliability limits determined by gate oxide thicknesses of thetransistors (e.g., V_(gs)<6 V).

For a voltage level of input DC signal V_(DD1) equal to 5 V, drains ofcascode transistor 404 and cascode transistor 406 will see voltagesslightly higher than 3×V_(DD1)=15 V. Cascode transistor 404 and cascodetransistor 406 enable fast restart of the oscillator by presenting asudden large voltage (e.g., a voltage above the latch crossover point,i.e., the point at which the gate-to-source voltage of latch transistor410 equals the gate-to-source voltage of latch transistor 408) acrosslatch transistor 408 and latch transistor 410. Voltages applied to latchtransistor 408 and latch transistor 410 are precisely controlled so thatthose transistors enter the triode mode of operation and turn off at anappropriate time with little or no crossover time (i.e., the transitiontime when latch transistor 408 and latch transistor 410 are conductingin the active mode of operation). Each of latch transistor 408 and latchtransistor 410 conducts during approximately one half of the cycle anddoes not conduct during the other half of the cycle. The capacitor ofoscillator circuit 402 can be fully differential (C_(p)), single-ended(C_(pa) and C_(pb)) or a combination of fully differential andsingle-ended. Every 2C units of capacitance on each single-ended branchis equivalent to C fully differential units. The total equivalentcapacitance seen by the oscillator circuit is C_(p) C_(p(a,b))/2.

Referring to FIG. 5, in at least one embodiment, oscillator circuit 402includes clamp transistor 412 and clamp transistor 414 coupled to thegate terminal of latch transistor 410 and the gate terminal of latchtransistor 408, respectively. Clamp transistor 412 and clamp transistor414 limit the gate-to-source voltages of latch transistor 408 and latchtransistor 410, respectively, to a maximum of approximatelyV_(DD1)|V_(tp)|. Clamp transistor 412 and clamp transistor 414 arep-type transistors configured to suppress any substantial couplingacross the drain-to-source parasitic overlap capacitance of cascodetransistor 404 and cascode transistor 406, respectively, if cascodetransistor 404 and cascode transistor 406 try to lift the gate-to-sourcevoltages of latch transistor 408 and latch transistor 410, respectively,above V_(DD1)+|V_(tp)|. For an exemplary 5 V CMOS process, V_(DD1) is 5V and |V_(tp)| is approximately 1 V, and the clamping occurs atapproximately 6 V, which is close to the maximum gate-to-source voltagethat a conventional 5 V transistor can withstand.

Referring to FIG. 6, in at least one embodiment, oscillator circuit 402includes clamp transistor 416 and clamp transistor 418 having sourceterminals coupled to the gate terminal of latch transistor 410 and thegate terminal of latch transistor 408, respectively. Clamp transistor416 and clamp transistor 418 each include a bulk terminal, a gateterminal, and a drain terminal that are coupled to the power supply nodeproviding input DC signal V_(DD1). Note that clamp transistor 416 andclamp transistor 418 are p-type transistors that have their n-type bulkterminal coupled to a corresponding drain node and not to acorresponding source node (i.e., a higher voltage node), as in typicalclamp transistor configurations. The configuration of clamp transistor416 and clamp transistor 418 results in two conduction paths for eachclamp transistor: a channel conduction path and a body diode conductionpath. Two conduction paths for each clamp transistor makes thisconfiguration faster than the typical clamp transistor configuration.Clamp transistor 416 and clamp transistor 418 limit the voltages of nodeV_(La) and node V_(Lb). If the voltages on node V_(La) or node V_(Lb)exceeds V_(DD1), then both the channel diode and the body diode of clamptransistor 416 and clamp transistor 418 start conducting, therebyclamping the gate voltages of latch transistor 410 and latch transistor408. Clamp transistor 416 and clamp transistor 418 are configured toreturn part of the clamped energy back to the power supply, therebyincreasing the efficiency of oscillator circuit 402.

The enable mechanism for controlling oscillator circuit 402 needs amechanism that reduces or eliminates excess energy that builds up in thetransformer coils upon restart and that can cause flying voltages on theterminals of the transformer (i.e., voltage levels much greater than3×V_(DD1) that develop on either node V_(Ha) or node V_(Hb) as a resultof releasing that excess energy to the capacitor(s) of the oscillator(e.g., C_(p), C_(pa), C_(pb)) as the oscillator restarts oscillating).Referring to FIGS. 7-9, embodiments of oscillator circuit 402 includesnubber circuit 420 and snubber circuit 422 coupled to the drainterminal of cascode transistor 404 and the drain terminal of cascodetransistor 406, respectively. Snubber circuit 420 and snubber circuit422 prevent the voltage on the drain terminal of cascode transistor 404and the voltage on the drain terminal of cascode transistor 406,respectively, from substantially exceeding 3×V_(DD1). As a result,snubber circuit 420 and snubber circuit 422 reduce or eliminate anynon-fundamental modes of oscillation (i.e., modes of oscillation with aneffective oscillation frequency other than

$\left. {f = \frac{1}{2\; \pi \sqrt{LC}}} \right)$

and force clean, well-bounded oscillation of oscillator circuit 402. Inaddition, snubber circuit 420 and snubber circuit 422 return at leastpart of the excess energy to the power supply. Snubber circuit 420 andsnubber circuit 422 may be sized to have a clamping voltage level justabove 3×V_(DD1). In at least one embodiment of oscillator circuit 402,snubber circuit 420 and snubber circuit 422 each include series-coupled,reverse-biased Zener diodes coupled in series with series-coupled,forward-biased diodes. Accordingly, the clamping voltage level equalsN₁×V_(Z)+N₂×V_(F), where N₁ and N₂ are integers greater than zero, V_(Z)is a knee voltage of the Zener diodes, and V_(F) is a forward voltage ofthe forward-biased diodes. Referring to FIG. 10, waveforms foroscillator circuit 402 illustrate pseudo-differential signals on nodesV_(Hb) and V_(Ha), with peak voltages slightly higher than 3×V_(DD1).

Referring to FIG. 2, oscillator circuit 202 converts input DC signalV_(DD1) to an AC signal (e.g., the pseudo-differential signal on nodesTX+ and TX−). Transformer 209 converts that AC signal into a second ACsignal (e.g., the pseudo-differential signal on nodes RX+ and RX−).AC/DC power converter circuit 204 receives the second AC signal fromconductive coil 208 and converts the second AC signal (e.g., thepseudo-differential signal on nodes RX+ and RX−) into output DC signalV_(DD2) that is electrically isolated from the input DC signal V_(DD1).AC/DC power converter circuit 204 includes a full-wave rectifiercircuit. Referring to FIGS. 2 and 12, in at least one embodiment, toimprove the efficiency of power transfer device 200 as compared toefficiency realized by conventional power transfer devices, conductivecoil 208 includes a center tap coupled to a ground node and rectifiercircuit 403 includes Schottky diode 1202 and Schottky diode 1204.

In general, a Schottky diode (i.e., hot carrier diode) is asemiconductor diode formed by a junction of a semiconductor with a metaland is characterized to have a fast switching speed and low voltagedrop. The Schottky diode can sustain high forward currents at lowervoltage drops than would exist in typical diffused pn-junction diodes.An exemplary Schottky diode forward voltage is approximately 150 mV-450mV, while a typical silicon diode has a forward voltage of approximately600 mV-700 mV. The lower forward voltage requirement improves systemefficiency. Typically, Schottky diodes are not available in conventionalCMOS manufacturing technologies because their manufacture requiresadditional mask layers and processing steps. However, Schottky diodesmay be available with conventional CMOS devices in an exemplarymixed-signal integrated circuit manufacturing process (e.g.,bipolar-CMOS-DMOS manufacturing process). Schottky diode 1202 andSchottky diode 1204 withstand voltages of greater than 10 V in a typicalapplication. The secondary-side half-windings alternate rectifying andadding charge to capacitor C1. Since only half of the transformerdelivers power to the output capacitor for a particular half-cycle, theoutput voltage that can be developed across C1 is limited. However, onlyone Schottky diode contributes to conduction losses according to whichpath is conducting at a particular time. Schottky diodes that have highcurrent density and relatively low reverse breakdown voltage may be usedto reduce area of the rectifier circuit. If Schottky diodes are notavailable, regular diodes may be used, but result in a lossier system.

Referring to FIG. 13, in at least one embodiment of a power transferdevice, rectifier circuit 403 includes conductive coil 208 with anunconnected center tap. Instead of a two-diode rectification structure,rectifier circuit 403 includes a four-diode rectification structure.Diode 1204, diode 1206, diode 1208, and diode 1210 are Schottky diodes,but regular diodes, which have higher losses across the diode, may beused. The embodiment of FIG. 13 allows a larger range of output voltagelevels for output DC signal V_(DD2) since the entirety of conductivecoil 208 delivers energy to the load during each half-cycle, as comparedto the embodiment of FIG. 12. However, the embodiment of FIG. 13 hasincreased conduction losses because the average output load currentconducts across two diodes and incurs two diode forward-voltage drops,instead of one diode forward-voltage drop of the embodiment of FIG. 12.

Replacing diode 1208 and diode 1206 of the embodiment of FIG. 13 withtransistors 1212 and 1214 of FIG. 14, improves efficiency of rectifiercircuit 403 by reducing conduction losses and the voltage drop acrosstransistor 1212 and transistor 1214 can be made lower than the forwardvoltage drop of the Schottky diodes, as compared to the embodiment ofFIG. 13. Referring to FIG. 14, in at least one embodiment, rectifiercircuit 403 includes Schottky diode 1202 and Schottky diode 1204integrated with conventional CMOS devices (e.g., cross-coupled n-typetransistor 1212 and n-type transistor 1214). Conductive coil 208 is notcoupled at a center tap. Regulating the output voltage level at anoutput DC signal V_(DD2) such that V_(DD2)+V_(F) falls below the maximumgate-to-source voltage of transistor 1212 and transistor 1214 alleviatesreliability concerns related to the maximum gate-to-source voltage oftransistor 1212 and transistor 1214. When either of transistor 1212 ortransistor 1214 of FIG. 14 conducts (e.g., the path through transistor1214, conductive coil 208, and diode 1202 or the path through transistor1212, conductive coil 208, and diode 1204), both the channel and thebody diode of transistor 1212 conduct, thus reducing conduction lossesas compared to the four-diode embodiment of FIG. 13. Referring to FIGS.14 and 15, waveform 1502 illustrates conduction through transistor 1212,conductive coil 208, and diode 1204 and waveform 1504 illustratesconduction through transistor 1214, conductive coil 208, and diode 1202.Waveform 1506 illustrates the rectified DC voltage (5 V) developed asthe output DC signal V_(DD2) across capacitor C1.

Referring to FIG. 16, power transfer device 1600 includes oscillatorcircuit 402 coupled to conductive coil 206, and is configured to deliverenergy from a first voltage domain across an isolation barrier viatransformer 209, which delivers power to a second voltage domain of load1650 via second conductive coil 208 and rectifier circuit 403.Typically, load 1650 and capacitor C1 are external to a package housingpower transfer device 1600. Oscillator circuit 402 is controlled with anenable/disable signal that controls the amount of energy delivered tothe secondary side of transformer 209, thereby regulating output DCsignal V_(DD2). Referring to FIGS. 16 and 17, power transfer device 1600isolates the first voltage domain on the primary side from the secondvoltage domain on the secondary side and allows data transfer betweenthe primary side and the secondary side. No external DC-to-DC converteris necessary to power up the second voltage domain. Transformer 209 isan air core transformer that is integrated in package 1702 (e.g., a widebody small outline integrated circuit (WBSOIC) package). Integration oftransformer 209 in power transfer device 1600 reduces bill of materialcosts and board space requirements of a voltage isolation system in anintended application (e.g., industrial and automotive applications).

FIGS. 17 and 18 illustrate a plan view of package 1702 housing powertransfer device 1600 and transformer 209 and a cross-sectional view ofcross-section 1800 of transformer 209 disposed on conductors in package1702. In at least one embodiment, transformer 209 is an air coretransformer with a 1:N turns ratio, where N can be approximately one.Each conductive coil of transformer 209 includes two turns each, and hasa planar spiral structure. However, one of skill in the art willappreciate that the teachings herein can be utilized with othertransformers using other turn ratios and/or other numbers of turns percoil. Transformer 209 converts the AC electrical energy on the primaryside into magnetic flux, which is coupled into the secondary side ofpower transfer device 1600. Transferring energy from the primary side tothe secondary side requires reinforced isolation, i.e., the primary andsecondary sides of transformer 209 need to be able to withstand voltagesurges greater than 5 kV RMS. Accordingly, the material used in the coreof the transformer that isolates conductive coil 206 from conductivecoil 208 needs to be able to withstand voltage surges greater than 5 kVRMS. Transformer 209 has a physical design and is formed using materialsthat reduce series resistance of conductive coil 206 and conductive coil208 and improve the quality factor of the transformer 209, whichincreases the efficiency of transformer 209.

In at least one embodiment, package 1702 houses power transfer device1600 and transformer 209 is formed using insulating substrate 1802.Insulating substrate 1802 is a glass substrate having a high transitiontemperature (i.e., a high Tg, e.g., Tg of at least approximately 150C)and a low dielectric constant (e.g., borosilicate glass, e.g., Tg ofapproximately 150), a resin-based substrate (e.g., Bismaleimide-Triazine(BT)), or a glass-reinforced epoxy laminate (FR-4). By formingtransformer 209 on insulating substrate 1802, transformer 209 can bedisposed directly on conductor 1705, conductor 1706, conductor 1707, andconductor 1708, which may be formed from plated copper or otherconductor within package 1702. Although insulating substrate 1802 isphysically in contact with conductor 1705, conductor 1706, conductor1707, and conductor 1708, transformer 209 is electrically isolated fromconductor 1705, conductor 1706, conductor 1707, and conductor 1708,thereby reducing physical size requirements for a package housingtransformer 209 in conjunction with other integrated circuits of powertransfer device 1600. For example, integrated circuit 1710 includesoscillator circuit 402, integrated circuit 1712 includes rectifiercircuit 403, integrated circuit 1718 includes feedback and faulttolerance circuitry, integrated circuit 1714 includes communicationchannel receiver circuitry, and integrated circuit 1716 includescommunication channel transmitter circuitry. However, in at least oneembodiment of power transfer device 1600, circuits of integrated circuit1712, integrated circuit 1716, and integrated circuit 1718 (e.g.,rectifier circuit 403, feedback and fault tolerant circuitry, andisolation channel transmitter circuitry) are integrated in fewerintegrated circuit die or a single integrated circuit die that are/iscoupled to transformer device 1704. Similarly, in at least oneembodiment of power transfer device 1600, circuits of integrated circuit1710 and integrated circuit 1714 (e.g., oscillator circuit 402 andcommunication channel receiver circuitry) are integrated in a singleintegrated circuit die that is coupled to transformer device 1704.

Integrated circuit 1710 and integrated circuit 1712 are coupled totransformer device 1704 using wire bonding or other integrated circuitinterconnect. In at least one embodiment, first conductive coil 206 isformed from a conductive layer followed by conventionalphotolithographic patterning. For example, a conductive layer (e.g., acopper layer) is formed on insulating substrate 1802. A photoresist isapplied and a reticle including a pattern for first conductive coil 206is used to selectively expose the photoresist material. Themanufacturing process removes unwanted material (e.g., unwanted materialis etched away). Instead of a subtractive patterning process, anadditive patterning process may be used to form conductive structuresonly in regions that need the material. Insulating layer 1804 is formedon first conductive coil 206. Then, second conductive coil 208 andinterconnection structures (not shown) are formed on insulating layer1804. Insulating layer 1804 may be any low dielectric constant materialhaving a high dielectric strength (e.g., epoxy-based photoresist, hightemperature polyimides, silicon dioxide or other thin film materialhaving a dielectric constant of less than 10). Thus, conductive coil 206and conductive coil 208 are formed in different layers on insulatingsubstrate 1802. Transformer 209 may include ground pins to increase heatdissipation and reduce junction temperature rise. Transformer 209 mayinclude two-turn conductive coils that have dimensions that reduceelectromagnetic interference (e.g., symmetrical coils with current flowin opposing directions) and achieve sufficient efficiency.

Referring to FIGS. 16 and 17, feedback circuit 1608 is a hystereticcircuit that regulates output DC signal V_(DD2) based on voltage V_(SNS)on a terminal SNS, which is a voltage-divided version of output DCsignal V_(DD2) generated by a resistor divider network. Hysteresis isused to generate a feedback signal that controls oscillator circuit 402to maintain the output voltage level of output DC signal V_(DD2) withina target voltage range. In an exemplary embodiment of power transferdevice 1600, V_(DD2),MAX is 5.02 V and V_(DD2),MIN is 4.98 V, resultingin an average regulated V_(DD2) value of 5.0 V. The resistor dividernetwork may be on-chip (e.g., integrated circuit 1712 or integratedcircuit 1718), off-chip (e.g., resistor R1 and resistor R2 areimplemented using discrete transistors in package 1702) and/or areexternal to package 1702 and coupled via one or more pins of package1702. In at least one embodiment of power transfer device 1600, only onepin (e.g., terminal SNS) is required to control output DC signal V_(DD2)and a hysteretic band of the voltage regulator of feedback circuit 1608,although in other embodiments, additional pins may be used.

Referring to FIGS. 16 and 19-21, feedback circuit 1608 includescomparator 1902, which compares voltage V_(SNS) on terminal SNS toreference voltage V_(REF) (e.g., a reference voltage generated bybandgap reference circuit 1612). When comparator 1902 detects thatvoltage V_(SNS) exceeds first threshold voltage V_(DD2),MAX, which isbased on reference voltage V_(REF), the output of comparator 1902changes output signal levels. In typical steady-state operation, (e.g.,TSHUT configures lockout circuit 1910 to pass a driven version of theoutput of comparator 1902), transmitter 1602 communicates the output ofcomparator 1902 from the secondary side to the primary side across theisolation barrier. The output of comparator 1902 may be converted to amodulated signal for transmission. For example, transmitter 1602 usesoscillator 1610 for on-off keying modulation to communicate a digitalrepresentation of the output of comparator 1902 across capacitivechannel 1606 to receiver 1604. The primary side generatesENABLE/DISABLE_B based on the received digital signal and controlsoscillator circuit 402 accordingly (e.g., disables cascode transistorsin oscillator circuit 402 to pause power transfer). When oscillatorcircuit 402 is disabled, load 1650 on the secondary side starts todischarge capacitor C1. As a result, the voltage level of output DCsignal V_(DD2) drops at a rate equal to

$\frac{{dVDD}\; 2}{dt} = {- {\frac{I_{load}}{C\; 1}.}}$

After the voltage level of output DC signal V_(DD2) crosses secondthreshold voltage V_(DD2),MIN, comparator 1902 changes the level of itsoutput signal. The change in voltage level is communicated from thesecondary side to the primary side across the isolation barrier. Thatchange in level causes the primary side to enable oscillator circuit402, which causes the voltage level of output DC signal V_(DD2) to rampup again. Output DC signal V_(DD2) may have a small AC ripple at twicethe oscillator frequency caused by the rectifier. That AC ripple ispresent only when the oscillator is on and when the voltage level ofoutput DC signal V_(DD2) is ramping up to first threshold voltageV_(DD2),MAX. An inherent delay of the received ON and OFF signalsgenerated by on-off keying signaling causes a small DC offset of outputDC signal V_(DD2) that may be reduced by reducing delay of the feedbackchannel.

Referring to FIGS. 16 and 19, the reference voltage V_(REF) and theratio of resistances of resistor R1 and resistor R2 determines thevoltage level of output DC signal V_(DD2) since

$V_{{DD}\; 2} = {V_{REF} \times {\frac{\left( {{R\; 1} + {R\; 2}} \right)}{R\; 2}.}}$

Hysteretic thresholds, first threshold voltage V_(DD2),MAX and secondthreshold voltage V_(DD2),MIN are programmed to target levels using acurrent I1 that is sourced by p-type transistor 1904 or sunk by n-typetransistor 1906 to/from the resistor network including resistor R1 andresistor R2:

$V_{{{DD}\mspace{11mu} 2},{MAX}} = {\frac{V_{ref}\left( {R_{1} + R_{2}} \right)}{R_{2}} + {I_{1}R_{1}}}$$V_{{{DD}\mspace{11mu} 2},{MIN}} = {\frac{V_{ref}\left( {R_{1} + R_{2}} \right)}{R_{2}} - {I_{1}{R_{1}.}}}$

Accordingly, a hysteretic band of the feedback signal is controlledindependently of the voltage level of output DC signal V_(DD2) by usinganalog techniques:V_(HYS)=V_(DD2,MAX)−V_(DD2,MIN)=2×I1×R1. Oscillator circuit 402 providesa fixed DC current to the secondary side and the load capacitor. Atsteady state, when the voltage level of output DC signal V_(DD2) movesbetween first threshold voltage V_(DD2,MAX) and second threshold voltageV_(DD2),MIN, capacitor C1 charges at a constant rate of approximately

$\frac{{dVDD}\; 2}{dt} = \frac{I_{out} - I_{load}}{C\; 1}$

and discharges at a constant rate of approximately

$\frac{{dVDD}\; 2}{dt} = {- {\frac{I_{load}}{C\; 1}.}}$

At steady-state,

$\frac{{dVDD}\; 2}{dt} = {V_{HYS}.}$

Therefore,

$t_{off} = {C\; 1 \times \frac{V_{HYS}}{I_{load}}}$

and the frequency of enabling and disabling of oscillator circuit 402 toachieve voltage regulation is

$\frac{1}{t_{on} + t_{off}},$

which is a function of C1, V_(HYS), and I_(load), and may vary accordingto particular manufacturing conditions. The frequency of feedbackchannel may be adjusted by selecting appropriate values for C1 andV_(HYS) for particular load conditions.

Referring to FIG. 20, in at least one embodiment of feedback circuit1608, rather than using current sources, resistor R3, is included inaddition to resistor R1 and resistor R2:

${V_{{{DD}\; 2},{MAX}} = \frac{\left( {{R_{1}R_{2}} + {R_{1}R_{3}} + {R_{2}R_{3}}} \right)V_{ref}}{R_{2}R_{3}}};{and}$$V_{{{DD}\mspace{11mu} 2},{MIN}} = {\frac{V_{ref}\left( {{R_{1}R_{2}} + {R_{1}R_{3}} + {R_{2}R_{3}}} \right)}{\left( {R_{1} + R_{3}} \right)R_{2}}.}$

Contrary to the embodiment described above where the average voltagelevel of output DC signal V_(DD2) is defined by

$V_{REF} \times \frac{\left( {{R\; 1} + {R\; 2}} \right)}{R\; 2}$

with a symmetrical hysteresis band V_(HYS)=2×I1×R1 evenly distributedaround the average voltage level of output DC signal V_(DD2), the upperand lower hysteresis thresholds of the embodiment of FIG. 20 are definedby more elaborate equations that are more complex to calculate.Nevertheless, the circuit provides the target upper and lower hystereticthresholds using a simpler implementation. Resistor R3 may be includedinternally to the integrated circuit while resistor R1 and resistor R2remain external to allow programmability of the hysteretic band andoutput voltage level using a single pin of the device by selection offirst resistance of resistor R1 and second resistance of resistor R2.

Referring to FIG. 16, in at least one embodiment of power transferdevice 1600, thermal shutdown circuit 1618 and thermal shutdown circuit1620 protect power transfer device 1600 from over-temperature events,which may occur due to ambient temperature being out of a specifiedrange, or due to a fault in power transfer device 1600. Thermal shutdowncircuit 1618 generates a primary-side thermal shutdown control signal inresponse to the temperature on a primary side integrated circuitexceeding a predetermined junction temperature (e.g., 150 C). Theprimary-side thermal shutdown control signal may be used bytimer/oscillator enable circuit 1622 to disable oscillator circuit 402,which is the predominate source of power dissipation of power transferdevice 1600. While oscillator circuit 402 is disabled, the junctiontemperature of power transfer device 1600 decreases, thereby protectingdevices of the primary side. Similarly, on the secondary side, thermalshutdown circuit 1620 generates a corresponding thermal shutdown controlsignal in response to the temperature on the secondary side integratedcircuit exceeding a predetermined junction temperature (e.g., 150 C). Asecondary-side thermal shutdown control signal may be used by feedbackcircuit 1608 to provide a feedback signal that, when transmitted acrossthe isolation channel, disables oscillator circuit 402, thereby allowingthe junction temperature of power transfer device 1600 to decrease andprotect devices on the secondary side. The primary side and thesecondary side can cause a thermal shutdown of oscillator circuit 402independently.

Referring to FIG. 16, in at least one embodiment of power transferdevice 1600, undervoltage lockout circuit 1614 and undervoltage lockoutcircuit 1616 reduce or eliminate erroneous operation during devicestartup and device shutdown, or when input DC signal V_(DD1) has a levelbelow its specified operating range. The primary side and secondary sidecan cause power transfer device 1600 to enter or exit an undervoltagelockout state independently. Undervoltage lockout circuit 1614 preventsfalse turn-on or false turn-off of oscillator circuit 402. Undervoltagelockout circuit 1614 generates a gating signal using a voltage detectorthat detects if the voltage level of input DC signal V_(DD1) is stableand has crossed a predetermined threshold voltage. Gating logic on theprimary side generates gating signal VDD1_OK, which is used by thermalshutdown circuit 1618 to configure circuitry to reduce or eliminateexcessive transient currents when the power supply is still coming upand has not yet stabilized. Gating all logic on the primary side withgating signal VDD1_OK reduces or eliminates incorrect decisions made inthe control path due to low supply voltage from turning off oscillatorcircuit 402. Similarly, on the secondary side, undervoltage lockoutcircuit 1616 monitors output DC signal V_(DD2). If the voltage level ofoutput DC signal V_(DD2) crosses a predetermined threshold voltage,undervoltage lockout circuit 1616 generates control signal VDD2_OK thatis used by thermal shutdown circuit 1620 to enable the output offeedback circuit 1608 to regulate the voltage level of output DC signalV_(DD2).

In at least one embodiment, power transfer device 1600, oscillatorcircuit 402 includes the LC tank-based oscillator having a cross-coupledn-type transistor latch, as described above, which limits the peakcurrent that can flow through transformer 209. When the secondary sideis shorted to ground due to a fault, the peak current limitation ofoscillator circuit 402 reduces or eliminates excessive current draw froman input node providing input DC signal V_(DD1). Those limits on thepeak current are described as follows:

${{Ipeak}_{primary} = \frac{2 \times V_{{DD}\; 1}}{{2\; \pi \; f_{0}{L\left( {1 - k^{2}} \right)}} + R_{s}}};{and}$${{Ipeak}_{secondary} = {k \times \frac{2 \times V_{{DD}\; 1}}{{2\; \pi \; f_{0}{L\left( {1 - k^{2}} \right)}} + R_{s}}}},$

where k is the mutual inductance of the transformer, and R_(S) is theequivalent series resistance of the primary winding. In an exemplaryembodiment, the inductance of conductive coil 206, L=100 nH, thefundamental frequency of oscillator circuit 402, f₀=75 MHz, and k=0.6,and R_(S)=1.4Ω. Accordingly, Ipeak_(primary) is approximately 316 mA andIpeak_(secondary) is approximately 200 mA.

In at least one embodiment, power transfer device 1600, includestimer/oscillator enable circuit 1622 on the primary side.Timer/oscillator enable circuit 1622 improves fault tolerance of powertransfer device 1600 in response to malfunctioning of the feedbackchannel. The feedback channel may be inoperative in response to a faultcondition on the secondary side or if the load is pulled to ground via asmall, finite resistance causing the oscillator circuit 402 tocontinuously transfer power to the secondary side. The continuoustransfer of power from the primary side to the secondary side couldcause heating of a secondary side integrated circuit that impactsreliability of the secondary side integrated circuit. For example,excessive junction heating on the secondary side causes thermal shutdownof the secondary side, but transmitter 1602 may be unable to transmitthe shutdown signal to the primary side due to a common-mode transientevent or fault condition. To reduce or eliminate overstress of deviceson the secondary side, timer/oscillator enable circuit 1622 monitors thereceived feedback signal for a predetermined period of time (e.g., 10ms). In at least one embodiment, timer/oscillator enable circuit 1622includes a counter that counts the number of transitions of the receivedenable/disable feedback control signal provided by receiver 1604 andcompares the count to a predetermined threshold count, after thepredetermined period. If an insufficient number of transitions of thereceived feedback signal level occur during that period,timer/oscillator enable circuit 1622 disables oscillator circuit 402 fora second predetermined period (e.g., 10 ms). After expiration of thesecond predetermined period, timer/oscillator enable circuit 1622resets. Timer/oscillator enable circuit 1622 continues to monitor andrespond any insufficient number of transitions of the received feedbacksignal until the fault condition disappears and the communicationschannel and received feedback signal become active again.

In at least one embodiment, power transfer device 1600, includesovervoltage protection circuit 1624 on the secondary side to reduce oreliminate driving a load with a voltage level of output DC signalV_(DD2) that exceeds reliability specifications. For example, a faultmay cause the communications channel to malfunction and not update thereceived feedback signal, which causes oscillator circuit 402 totransfer power for a longer period than necessary to charge capacitor C1to a voltage level corresponding to first threshold voltage V_(DD2,MAX)and thereby causes the voltage level of output DC signal V_(DD2) toexceed the voltage level corresponding to first threshold voltageV_(DD2,MAX) (e.g., voltages of 9 V for a 5 V V_(DD2,MAX)). Suchexcessive voltage on the secondary side could damage a device in theload. To reduce or eliminate the likelihood of excessive levels ofoutput DC signal V_(DD2), overvoltage protection circuit 1624 draws anyexcess current on the secondary side and sinks that excess current toground, thereby clamping the voltage level of output DC signal V_(DD2)and preventing it from further rise. In at least one embodiment,overvoltage protection circuit 1624 includes an active shunt regulatorthat is configured as an active clamp. An exemplary shut regulator isimplemented using feedback circuit techniques that create a scaledversion of reference voltage V_(REF) (e.g., 1.1×V_(REF)) and comparesthat scaled version of reference voltage V_(REF) to voltage V_(SNS)using an error amplifier. The output of the error amplifier activates aclamping device if voltage V_(SNS) exceeds the scaled version ofreference voltage V_(REF). That clamping device shunts the excesscurrent to ground and regulates voltage V_(SNS) to be approximatelyequal to the voltage level of the scaled version of reference voltageV_(REF), thereby preventing the voltage level of output DC signalV_(DD2) from rising further. The scaled version of reference voltageV_(REF) sets the active clamping level to be approximately 10% above thenominal voltage level of output DC signal V_(DD2), which is within thereliability limits of an external load. For example, if the targetvoltage level for output DC signal V_(DD2) is 5V, the active clampengages when the voltage level of output DC signal V_(DD2) exceeds 5.5V.During normal operation (i.e., a no-fault mode of operation), voltageV_(SNS) is less than 1.1×V_(REF) and the shunt device is inactive inthat state. The active clamp may be disabled while the voltage level ofoutput DC signal V_(DD2) ramps up (e.g., during a power-up sequence) andmay only be enabled when the voltage level of output DC signal V_(DD2)is close to its regulated voltage level and VDD2_OK indicates no faultcondition.

While circuits and physical structures have been generally presumed indescribing embodiments of the invention, it is well recognized that inmodern semiconductor design and fabrication, physical structures andcircuits may be embodied in computer-readable descriptive form suitablefor use in subsequent design, simulation, test or fabrication stages.Structures and functionality presented as discrete components in theexemplary configurations may be implemented as a combined structure orcomponent. Various embodiments of the invention are contemplated toinclude circuits, systems of circuits, related methods, and tangiblecomputer-readable medium having encodings thereon (e.g., VHSIC HardwareDescription Language (VHDL), Verilog, GDSII data, Electronic DesignInterchange Format (EDIF), and/or Gerber file) of such circuits,systems, and methods, all as described herein, and as defined in theappended claims. In addition, the computer-readable media may storeinstructions as well as data that can be used to implement theinvention. The instructions/data may be related to hardware, software,firmware or combinations thereof.

Thus, a power transfer device having an integrated transformer, arelatively small size, high efficiency, and with built-in faulttolerance, and programmable output voltage, voltage ripple, andfrequency of DC/AC power conversion has been described. The descriptionof the invention set forth herein is illustrative and is not intended tolimit the scope of the invention as set forth in the following claims.For example, while the invention has been described in embodiments of apower transfer device, techniques described herein may be combined withother isolation products, e.g., digital isolators, analog isolators, andgate drivers in the same package. Variations and modifications of theembodiments disclosed herein, may be made based on the description setforth herein, without departing from the scope of the invention as setforth in the following claims.

What is claimed is:
 1. A method for operating a power transfer device,the method comprising: converting an input DC signal into an output DCsignal, the output DC signal being electrically isolated from the inputDC signal, wherein the converting comprises: biasing an oscillatorcircuit with the input DC signal; and developing a pseudo-differentialsignal on a first node of the oscillator circuit and a second node ofthe oscillator circuit, the pseudo-differential signal having a peakvoltage of at least three times an input voltage level of the input DCsignal.
 2. The method, as recited in claim 1, wherein transistors in alatch circuit of the oscillator circuit each have a breakdown voltagethat is just over the input voltage level.
 3. The method, as recited inclaim 2, wherein the converting further comprises: clampinggate-to-source voltages of the transistors in the latch circuit to amaximum voltage of the input voltage level.
 4. The method, as recited inclaim 1, wherein the converting further comprises: generating a first ACsignal by selectively enabling the oscillator circuit according to acontrol signal.
 5. The method, as recited in claim 4, wherein thecontrol signal is based on a received feedback signal and a statusindicator, the received feedback signal being received using acommunications channel and the status indicator indicating an operationstatus of the communications channel across an isolation barrier.
 6. Themethod, as recited in claim 5, wherein the converting further comprises:generating the status indicator based on a change in signal level of thereceived feedback signal provided by the communications channel within apredetermined period.
 7. The method, as recited in claim 6, furthercomprising: disabling the oscillator circuit in response to the statusindicator indicating a lack of change in the signal level of thereceived feedback signal within the predetermined period.
 8. The method,as recited in claim 4, wherein the converting further comprises:converting the first AC signal into a second AC signal, the first ACsignal being electrically isolated from the second AC signal; generatingthe output DC signal by rectifying the second AC signal, the output DCsignal being electrically isolated from the input DC signal; generatinga feedback signal based on the output DC signal; and communicating thefeedback signal across an isolation barrier using a communicationschannel.
 9. The method, as recited in claim 1, wherein the convertingfurther comprises: limiting voltages on drain terminals of cascodedevices in the oscillator circuit to be at most, just over three timesthe input voltage level, thereby snubbing non-fundamental modes ofoscillation of the oscillator circuit and returning excess energy fromthe cascode devices to an input power supply node receiving the input DCsignal.
 10. A method for manufacturing a power transfer device, themethod comprising: forming a primary-side circuit on a first integratedcircuit die configured to receive an input DC signal; forming asecondary-side circuit on a second integrated circuit die configured toprovide an output DC signal; forming a transformer on an insulatingsubstrate; electrically coupling the transformer between theprimary-side circuit and the secondary-side circuit; and disposing theinsulating substrate directly on conductors in a package housing thefirst integrated circuit die and the second integrated circuit die, thetransformer being electrically isolated from the conductors.
 11. Themethod, recited in claim 10, wherein the primary-side circuit comprises:an input power supply node configured to receive the input DC signal;and an oscillator circuit configured as a power amplifier of a DC/ACpower converter responsive to the input DC signal and an oscillatorenable signal to provide a first AC signal; and a circuit configured togenerate the oscillator enable signal based on a received feedbacksignal and a status indicator indicating an operation status of acommunications channel across an isolation barrier, the receivedfeedback signal being received via the communications channel.
 12. Themethod, recited in claim 11, wherein the transformer is configured toconvert the first AC signal to a second AC signal.
 13. The method,recited in claim 12, wherein the secondary-side circuit is electricallyisolated from the primary-side circuit, and is configured to convert thesecond AC signal to the output DC signal.
 14. The method, recited inclaim 13, wherein the secondary-side circuit comprises: an output powersupply node; and a second circuit configured to generate a feedbacksignal based on a predetermined reference voltage and the output DCsignal on the output power supply node, wherein the secondary-sidecircuit further comprises an isolation channel transmitter configured totransmit the feedback signal across the isolation barrier using thecommunications channel, and wherein the primary-side circuit furthercomprises an isolation channel receiver configured to receive thefeedback signal from the secondary-side circuit via the communicationschannel.
 15. The method, recited in claim 10, further comprising:coupling at least one resistor to a terminal of the package, wherein alevel of the output DC signal is based on a resistance of the at leastone resistor.
 16. The power transfer device manufactured by the methodof claim
 10. 17. A power transfer device having a primary side and asecondary side isolated from the primary side by an isolation barrier,the power transfer device comprising: a first power supply node; asecond power supply node; and an oscillator circuit coupled to the firstpower supply node and the second power supply node, the oscillatorcircuit comprising: a primary-side conductive coil having a center tapcoupled to the first power supply node; wherein the oscillator circuitis configured as a DC/AC power converter circuit that receives, on thefirst power supply node, an input DC signal, and regulates an outputvoltage of the secondary side based on a feedback signal received fromthe secondary side via a communication channel across the isolationbarrier.
 18. The power transfer device, recited in claim 17, wherein theoscillator circuit comprises: a snubbing circuit configured to suppressnon-fundamental modes of oscillation of the oscillator circuit andreturn excess energy from the oscillator circuit to the first powersupply node.
 19. The power transfer device, recited in claim 17, whereinthe oscillator circuit further comprises: a latch circuit including apair of cross-coupled transistors each having a breakdown voltage thatis just over a voltage level of the input DC signal; and a cascodecircuit coupled to the pair of cross-coupled transistors and theprimary-side conductive coil.
 20. The power transfer device, recited inclaim 19, further comprising: a clamping circuit coupled between thesecond power supply node and a node between the latch circuit and thecascode circuit.